A profound need exists in many trades, disciplines, and industries for a truly accurate non-contact distance-measuring system. For example, the construction trades would benefit from a handheld device that emitted a visible laser beam that when aimed at a target, would measure the distance from the device to the target in less than one second, with accuracy better than one-sixteenth of an inch, independent of ambient temperature and other environmental constraints, and do so with readily available and inexpensive electronic components. To reach a wide audience and range of applicability, the distance measuring technology should be amenable to implementations incorporating infrared, microwave, millimeter-wave, or other electromagnetic wave carriers, as well as acoustic. Regardless of the carrier chosen, safety requirements, in particular eye safety, must be addressed.
There are four predominant electro-optical methods for active distance measurement: interferometric, triangulation, pulsed time-of-flight (TOF), and phase measuring. Interferometric methods result in accuracies of less than one micrometer over ranges of up to several millimeters. Triangulation techniques result in devices with accuracy in the micrometer range, but can only measure distances out to several inches. Devices based upon pulsed TOF and phase-measuring equipments, as well as equipments based upon a heretofore obscure technology called coherent burst, measure distances and velocities of similar magnitude to the present invention, but of significantly lesser accuracy and greater cost. Each of these are described below.
Pulsed TOF
In this method, a pulse of light, usually emitted from a laser, is transmitted to a target, and a portion of the light pulse reflected from the target is collected at the source location. The round trip transit time of the light pulse is measured, and the distance from the rangefinder to the target is d=ct/2, where d is the distance, c is the speed of light, t is the round trip transit time, and the factor of two accounts for the distance having to be traversed two times by the light pulse.
The pulsed TOF art is quite rich. An illustrative embodiment is depicted in FIG. 1, which is a block diagram simplification of a distance measuring circuitry 10. In this circuit, a start command input and a pulse generator 12 are coupled to the inputs to a transmitter 14. An output of the transmitter 14 is coupled to the input of a driver 16 and an input to the constant current source 24. An output of the driver 16 is coupled to the transmitting element 18. A receiving element 20 is positioned to receive light reflected back from target T. An output from the receiving element 20 is coupled to receiver 22 which has an output coupled to an input of the constant current source 24 and an input to a delay 26. An output of the delay 26 is coupled to an input to an analog-to-digital (A/D) converter 28 which has an output coupled to a digital processor. A capacitor 29 is coupled in parallel to a reset switch 30 and one terminal of the capacitor 29 and one terminal of the reset switch 30 is coupled to the constant current source 24 and the A/D converter 28.
In FIG. 1, the circuitry 10 has the pulse generator 12 that generates pulses that are gated by a start command at the AND gate 14 and passed to a laser driver 16 that is used to drive an emitter 18 of electromagnetic energy radiation for the duration of each of the pulses generated by the pulse generator 12. These short pulses of electromagnetic energy are directed to the remote target T, and a portion of the electromagnetic energy is reflected from the target T and received by a receiving element 20. Nominally the transmitting element 18 and the receiving element 20 are substantially collocated. The receiver 22 processes and amplifies the received pulse signal, and forwards the pulse signal onto the constant current source circuit 24.
The constant current source 24 also has as an input the transmit pulse signal output by the AND gate 14. The constant current source 24 creates an output signal that is a constant current, the start of the constant current beginning with the rising edge of the output of the AND gate 14, and ends with the rising edge of the pulse output by the receiver 22. Therefore, the duration of the constant current signal is proportional to the round trip travel time of the pulse energy transmitted to the remote target T by the transmitting element 18 and received by the receiving element 20, after subtracting off the electronic delays of the associated circuitry.
Next, the output of the constant current source 24 is fed to an integrating capacitor 29 which accumulates a charge and produces a voltage ramp at its output terminal. With this charging method, the capacitor 29 obtains a voltage at the end of the constant current signal output by the constant current source 24 that is proportional to the distance from the transmitting element 18, to the target T, and back to the receiving element 20. The voltage on capacitor 29 is directed an analog to digital (A/D) converter for downstream computer processing. The output of the receiver 22 is routed through a delay circuit 26, whose output in turn is routed to the start conversion input of the A/D converter 28. This signal causes an A/D conversion to commence well after the voltage on capacitor 29 has plateaued.
There are variations on the basic pulse TOF architecture. For example, one type of architecture teaches how the capacitor voltage can be downward sloping as the capacitor is discharged with a constant current source between the start and stop pulses. Instead of generating a voltage ramp, another type of architecture describes how a high-speed digital counter can be continuously incremented with a high frequency clocking signal after the start pulse occurs, and then terminates when the stop pulse occurs. This eliminates the need for an A/D converter as the output of the counter is already in a digital format. However, this counter approach has quantization errors, which is remedied by random dithering or interpolation methods. The counter or pulse TOF methods can be used for coarse range estimates, while phase measuring TOF, discussed below, is used for more precise range estimates. Alternately, a series of N pulses could be transmitted, in which a subsequent pulse is transmitted after the previous one is received, and the total time for these N pulses to be sent and received is measured. Thereafter, the time is divided by N to obtain a more precise estimate of a round trip transit time. A pulse train of a predetermined timing sequence could be used. An electronic correlation function is used to compare the delayed transmit sequence to the received sequence, and when correlation is found the delay has the round trip transit time of the pulse sequence.
All of these pulsed TOF methods are conceptually simple, although their implementation is usually complex and expensive. Specifically, to obtain an accurate distance estimate, the pulses must either be extremely short, or as is usually the case, must have fast low-high and high-low transitions. To obtain accuracies on the order of 0.1″, electronic bandwidths on the order of 1.0 gigahertz, or greater, are required in the transmission electronics, including the laser, as well as in the receive electronics, including the photodiode. Such broadband electronic components are expensive, and drive up the overall cost of the system. Furthermore, the distance signal processing is a two-stage affair. First, the distance information is encoded into a capacitor's voltage, and then secondly this voltage is converted into digital format for subsequent processing. A circuit that offers a single stage of processing is likely to be simpler, lower cost, and less error prone than a multi-stage system.
It is also difficult for pulse-TOF systems to measure velocity. One method is to transmit a series of pulses, and then analyze the Doppler shift of the returned pulses. A second way is to compute the velocity from the time rate of change of the estimated distance. Both methods suffer from accuracy problems, and are relatively expensive to implement because they require the use of broadband electronics to ensure the fidelity of the pulses.
However, pulsed TOF systems do offer one important advantage that is retained in the present invention: owing to their low duty cycle emission waveform they can be made eye-safe with visible light emissions, yet can also measure distances well beyond ten meters.
Phase Measuring
In phase measuring rangefinding, a periodic modulation signal, usually a sinusoid, is transmitted to the target, and an echo is received and amplified. The phase of the received signal will be delayed when compared to the phase of the transmitted signal because of the round trip transit time of the signal. The phase difference between the two signals is directly proportional to the distance to the target, according to the expression d=φλ/4π, where d is the distance from the rangefinder to the target, and λ is the wavelength of the modulating sinusoid (e.g., is 15 meters for a 20 MHz signal), and φ is the phase difference in radians. A range ambiguity arises every λ/2 meters of distance, in which the phase of the modulating signal is identical every Nλ/2 meters. This is one of the major drawbacks of phase measuring methods.
Another drawback is that since the modulation occurs in a continuous-wave fashion, the average power of the carrier must be high in order to be able to obtain a significant received signal for large target distances. High average carrier powers in the visible spectrum are not eye-safe. Yet another drawback concerns undesirable phase delay changes of the electronic circuitry with changes in ambient environmental conditions, especially temperature. Also, gain changes in AGC (Automatic-gain-control) circuitry will cause changes in phase as well, and these changes cannot be reliably calibrated and subtracted out with commonly used on-board reference methods.
For all these disadvantages, phase measuring rangefinders offer better intrinsic accuracy and lower cost than their pulsed TOF counterparts, arising primarily from their narrowband electronics.
There are two primary technologies used in phase measuring rangefinders: homodyne and heterodyne. Each of these are discussed below:
Heterodyne
A heterodyne demodulator is one in which a high frequency signal is mixed with a signal of a different frequency, and the resulting signal has components of the sum and the difference of the two frequencies. Typically the frequency difference between the two mixed signals is a constant known frequency. The resulting higher frequency, corresponding to the sum of the frequencies is usually ignored and removed through filtering. The lower frequency signal is amplified in a bandpass amplifier resulting in a signal that has a good signal to noise ratio owing to the fact that all out of band noise is filtered by the bandpass amplifier. This amplified signal is mixed yet again with another signal, this time having the same frequency, and low pass filtered, resulting in a low-noise DC component whose amplitude is proportional to the phase of the received signal. Alternately, if the target is moving, the DC signal will not be present, but instead a low frequency AC signal will be present, and the frequency of this signal is proportional to the velocity of the target because of the Doppler shift.
A functional block diagram of a heterodyning phase-measuring rangefinder 40 is shown in FIG. 2. A clock 42 is coupled to the inputs of a frequency synthesizer 44, a frequency synthesizer 56, and a divider 60. An output of the frequency synthesizer 44 is coupled to the input of the laser driver 46. An output of the frequency synthesizer 56 is coupled to the input of the mixer 54 and an output of the divider circuit 60 is coupled to the input of the mixer 62. The output of the laser driver 46 is coupled to the input of a laser diode 48. A photodiode 50 is positioned to receive light reflected from the target T. An output of the photodiode 50 is coupled to an input of a preamplifier 52 which has an output coupled to the mixer 54. An amplifier 58 is coupled between the mixer 54 and the mixer 62 and an output of mixer 62 is coupled to an input of an A/D converter 64 which has an output coupled to a digital processor.
In FIG. 2, a 3.64 MHz Clock 42 provides a reference frequency. A first frequency synthesizer 44 up converts this 3.64 MHz clock frequency to a frequency 455 kHz above 93 MHz. This sinusoidal frequency is then forwarded to a laser diode driver 46 which creates an electronic drive signal having the correct power and voltage characteristics required by the laser diode 48 so that the laser diode 48 emits an optical signal, the modulated light out. This modulated light out is directed to the remote target T by the operator, and a portion of the reflected modulated light is reflected back to the device by the remote target T. The phase of the reflected modulated light at the receiving photodiode 50 is different than the phase of the transmitted modulated light at the laser diode 48 owing to the round-trip propagation time delay of the optical signal. Further, the magnitude of the phase change is directly proportional to the distance between the remote target T and the rangefinder 40, provided the laser diode 48 and the photodiode 50 are substantially collocated. The photodiode 50 converts the received reflected modulated light optical signal into an electrical signal, and this weak electronic signal is amplified by a preamplifier 52.
The signal output by the preamp 52 is input to a first mixer 54 which has as a second input a 93.0 MHz signal generated by a second frequency synthesizer 56. The result of the mixing operation within the first mixer 54 is a 0.455 MHz signal. This frequency is nominally constant and is low enough to be economically filtered in the intermediate-frequency (IF) amplifier stage. That is, the IF amplifier 58 will amplify the 0.455 MHz signal and filters out all other signals not at the 0.455 MHz passband and thereby improves the signal to noise ratio of the 0.455 MHz signal at the output of amplifier 58. This filtered signal from amplifier 58 is used as an input to a second mixer 62 that has as a second input a 0.455 MHz signal generated by a divider circuit 60. The divider circuit 60 simply divides the 3.64 MHz signal generated by 3.64 MHz clock by a factor of eight to generate the 0.455 MHz signal. Since the two signals being mixed together within the second mixer 62 have the same frequency, the output signal will be a DC voltage whose amplitude is proportional to the cosine of the phase difference between the two signals input to the second mixer 62, which in turn means that the target distance is also proportional to the cosine of the DC signal. This DC signal is then input to an A/D converter 64 which converts the DC signal to a digital format for further digital processing.
While conceptually simple, this analog method of distance measurement is intrinsically incapable of measuring phase to the picosecond level needed for sub-millimeter accuracy. This arises primarily from non-linearities within the mixers and amplifiers, and from drift in gain and electronic signal delay times due to changes in ambient environmental conditions. Furthermore, since the DC signal is proportional to the cosine of the phase difference, there will be certain phase differences that result in imprecise phase estimation owing to the slow rate of change of the cosine function, most noticeably at nπ phase differences, where n is an integer. When the transmitted modulated light has a modulation frequency of 93.455 MHz, the wavelength of the modulation is λ=3×108/93,455,000=3.210 meters, and the distances of poor precision are centered around 3.210n/4=0.8025n meters. In other words, whenever the distance to the target is approximately 802.5n millimeters, the precision of the distance measurement is questionable.
There is yet another problem with the signal output from the mixer. Not only is it proportional to the cosine of the phase difference, but it is also a function of the amplitude of each of the signals being mixed together. If one of the signals amplitude unexpectedly changes due to noise or fluctuations in the return signal, then the interpretation of the mixer's output can lead to serious distance estimate inaccuracies.
Homodyne
A similar demodulation method utilizes homodyne electronic processing, in which the received signal is mixed with a signal having the same frequency. This is different than the heterodyne system described above where the received signal is first mixed with a signal having a different frequency. The result of homodyne mixing is that the first mixing stage results directly in the phase or low frequency AC signal for distance or velocity estimation. The second heterodyne mixing is eliminated, meaning less electronic components are utilized which translates into a cost savings, but typically the SNR is somewhat poorer than heterodyne-based distance and velocity measurement.
The homodyne phase measuring rangefinder has the same drawbacks of the heterodyning rangefinder, especially as related to nonlinearities within the electronic functions, particularly the phase splitter and the mixers, as well as the imprecision at distances proportional to nπ phase difference, and gain and delay drifts with changes in environmental conditions. Their mixer's outputs are also a function of the input signal amplitudes, and suffer from the same problems as discussed previously.
Both the homodyne and heterodyne methods suffer from the aforementioned range ambiguity problem, which can be remedied by utilizing a second, lower, operating frequency whose first ambiguity is beyond the operating range of the device. Lastly, since the laser is amplitude modulated for a relatively long period of time, and since the average laser output power must be limited to 1 mW for eye-safety considerations, the maximum range for visible laser emissions is therefore limited as well.
Other phase measuring include a phase measuring distance measuring system that uses light as the modulation carrier. A homodyne mixer can be used for electronic signal processing, while still incorporating an optical modulation carrier. Multiple modulation frequencies can be used to resolve the ambiguity problem and to improve the accuracy of the distance estimate. Heterodyne electronic signal processing methods can also be used in conjunction with two or more modulation frequencies.
Coherent Burst
Coherent burst technology is a significant improvement over the phase measuring and pulse-TOF distance measuring methods. Specifically, the coherent burst modulation waveform allows the maximum range to be increased without compromising eye safety, and since the modulation is bandlimited the resulting low cost circuitry and measurement accuracy is similar to that of the phase measuring methods. Coherent burst technology accomplishes this by combining the best of the phase-measuring and pulse-TOF methods, wherein a short series of bursts of amplitude modulated light is transmitted to the target. FIG. 4 illustrates the envelope of the coherent burst emission waveform, and FIG. 5 presents a magnified, and abbreviated, diagram of the coherent burst emission. The short bursts have pulse-like properties, in that they have a starting edge and a trailing edge, and a burst transmission can be used to start a counter or voltage ramp, and its reception from the target can be used to stop the counter or the voltage ramp, as described in the pulse TOF prior art discussion, above. This method can be used to provide a coarse estimate of the range, and therefore resolve the range ambiguity problem associated with phase measuring methods.
The coherent burst, being a short duration burst of amplitude modulated light, will also work with phase measuring methods, provided that the electronics comprising these phase measuring methods can respond and settle within the duration of a burst. Increasing the amplitude modulation frequency of a burst allows for increased measurement accuracy. Furthermore, by spacing the coherent bursts in time, high burst powers can be realized while maintaining an eye-safe average power, and long distances can be measured.
An illustrative functional diagram for a conventional embodiment of the coherent burst distance measuring method is presented in FIG. 3. A pulse generator 72 and a master oscillator 74 are coupled to inputs of an AND gate 76. An output of the AND gate is coupled to an input to a laser driver 78 which has an output coupled to an input to a laser diode 80. An output of a photo diode 82 is coupled to an input to a preamplifier 84. An output of preamplifier 84 is coupled to an input of bandpass filter and amplifier 86. An output of bandpass filter and amplifier 86 is coupled to inputs of multipliers 90 and 92. A phase shifter 88 has an input coupled to an output of master oscillator 74 and has outputs coupled to multipliers 90 and 92. Outputs of multipliers 90 and 92 and an output of pulse generator 72 are coupled to inputs of processor circuit 94.
In FIG. 3, a master oscillator 74 oscillates at the burst frequency, and whose signal is gated by an AND gate 76 by the pulses output of a pulse generator. The pulses correspond to the bursts, and the signal generated by the master oscillator 74 corresponds to the amplitude modulation within the burst. The composite burst signal is then input to a laser driver 78 which creates electronic signals suitable for driving a laser diode 80 in accordance with burst signal. The laser light is then directed onto a target T by the operator, whereupon a portion of it is reflected back by the target T to the photodiode 82 which is substantially collocated with, but offset from, the laser diode 80. The photodiode 82 converts the received optical signal into an electronic signal, which is then amplified by a preamplifier 84. The output of the preamplifier 84 is then directed to a bandlimited filter-amplifier 86 which amplifies the signal further, but also rejects all out of band noise thereby improving the signal to noise ratio of the signal. This signal is then fed to two multipliers 90 and 92 for subsequent processing. The master oscillator 74 has a second output, synchronized with the first output, that is directed to a phase shifter 88. The phase shifter 88 then has as an output two signals that are in sync with the input signal, but whose outputs have a 90° phase difference between them. One of these output signals is then fed to the second input of the In-Phase multiplier 92 and the other output signal is fed to the second input of the Quadrature multiplier 90. As with the second mixer of FIG. 2, the output of both multipliers 90 and 92 are DC voltages, assuming a stationary target, because the input frequencies are the same for both. Multipliers 90 and 92 together constitute a quadrature phase detector and their outputs are directed to a processing circuit 94 which converts the first and second quadrature signals into a phase estimate. The pulse signal is also input to the processor circuit 94 for ambiguity resolution according to the pulse-TOF methods described earlier.
Doppler shift of a coherent burst waveform can be used for target velocity estimation. More recently, a homodyne coherent burst system with quadrature electronic signal processing can be used for velocity estimation. Digital signal processing methods can also use coherent burst velocity estimation based upon the Doppler shift.
While the coherent burst method seems to intrinsically solve many of the problems of eye-safe distance measurement, some problems still remain. For example, the non-linearities found in phase-measuring amplifiers and mixers still remain. Secondly, the multiplexer circuit which generates the two signals having 90° phase differences must generate these signals with exactly 90° phase differences between them. Any errors in these differences will cause an uncalibratable distance error. Thirdly, the output of the mixers are still a function of the amplitude of the input signals, and, lastly, uncalibrated phase delays are introduced when Automatic-gain-control methods are employed. This topic is addressed in detail, below. In sum, these errors make it very difficult to economically achieve accuracy better than 0.1″ in a compact or hand-held distance measuring unit.
AGC Considerations
Unlike pulsed TOF ranging technology where the electronic signals are nominally saturating, gain controlling means are required for phase measurement ranging to prevent the electronic signals from saturating, yet keep the signals large enough to be usable. For phase measurement ranging, AGC is particularly problematic because when the gain of an electronic amplification circuit is changed, an RC time constant or semiconductor junction delay time within that circuit is also usually changed, meaning the delay of the signal through that circuit is also changed. When dealing with energy velocities at the speed of light, phase changes below 1 ps are required to keep the system measurement accuracy better than 1 mm. Further, since it is unknown what the correct AGC gain should be a priori it is extremely difficult to accurately calibrate this AGC-induced variable delay and subtract it from the distance measurement. This is a subtle yet important defect in the prior art.
Several automatic-gain-controlling means exist. For example, an electronic means to compress high amplitude electronic signals, but yet not compress low amplitude signals which arise from distant or non-reflective targets have been described. Electronic circuits of this type commonly introduce phase variations as their electronic gain changes, and these variations will not be referenced out by a reference optical path because the referencing will typically be made at a gain setting different than that used for the actual distance measurement. A variable gain amplifier can also be used as an AGC, but this type of circuit will also generally introduce more than one ps of varying signal delay.
An alternate way to maintain a constant amplitude electronic signal is to control the power output of the emitting device, such as with a laser diode. The power of the electromagnetic wave carrier can also be controlled in non-laser devices to accomplish the same end. But these methods just move the phase error problem from the receive signal path to the transmitter signal path. When the power levels of the emissive devices are changed, their operating characteristics change, and they will generally introduce several picoseconds of signal delay change. Furthermore, these errors will not be calibrated because the calibration will occur at one level of emission, and the actual distance measurement operation will occur at a different level having a different phase delay. Also, if the transmitted signal is attenuated too much, the emission will not be visible on the target and pointing accuracy will suffer.
Two emissive devices, one emitting a high power for longer distances or less reflective targets, and the other emitting a lower power, can also reduce the need for gain control elsewhere in the system. But this has the obvious disadvantage of increased cost and size, and reduced reliability.
Yet another means to circumvent the AGC problem is to use a resistor ladder in the receive path in which the electronic signal, at several different amplitudes, is presented to an electronic switch. The appropriate signal level is then selected for further processing. However, this approach also tends to be complex and costly.
A variant of this method involves one or more receive amplification paths which are implemented in parallel, and the electronic signal created by the path with the optimal fixed gain is selected for further processing. This method also suffers from increased complexity and cost.
Two photodiodes, a near PD and a far PD, can be used with their downstream amplification means to reduce the amount of variable gain control needed. But doubling the number of receive paths obviously comes with increased cost and complexity, and reduced reliability. Most of these methods also suffer from serious electronic signal delay calibration problems.
There are also ways to implement an optical AGC, which will not introduce unwanted changes in phase with signal amplitude or changes in gain. It is well known to those skilled in the art that light from a distant source will be brought to a focus at the focal plane of a lens. As the source is brought closer to the lens, the focus spot blurs and becomes larger in size at the focal plane. If a light detector of finite size is placed at the focal point, all of the light reflecting from a distant target and collected by the lens will be incident upon it. However, as the target is brought closer to the lens, the blur spot overfills the detector, and substantially less light, as a percentage of that collected by the lens, is incident upon the light detector. Indeed, in a compact optical arrangement, optical AGC can provide three orders of magnitude of gain control. This is a passive open-loop mechanism, requiring no control components, and is therefore highly desirable for its simplicity, low cost, and reliability.
An alternate optical AGC provides a means to reduce the diameter, and hence the power, of the laser beam by mechanically placing different apertures in front of the laser. This method does successfully implement an optical AGC function, but suffers from the common problems of increased cost and complexity.
Lastly, partial blocking of the collection lens can work as an optical AGC, but it too has increased complexity, reduced reliability, and should not be relied upon in and of itself because of its limited dynamic range.